1. Field of the Invention
This invention relates to a signal amplification system and, more particularly, to a system and method which enables linear amplification of a signal.
2. Description of Related Art
An ideal power amplifier amplifies an input signal with no waveshape alteration. The ideal power amplifier is therefore characterized as having a transfer function (input signal vs. output signal) which is linear with no transfer function discontinuities. In practice, however, a power amplifier has a transfer function with nonlinear and xe2x80x9clinearxe2x80x9d regions. Whether the power amplifier is operating in a linear or nonlinear region depends on the amplitude of the input signal. For the power amplifier to achieve as near to linear operation as possible, the power amplifier is designed to operate within its linear region given the range of possible input signal amplitudes. If the input signal has an amplitude which causes the power amplifier to operate outside the linear region, the power amplifier introduces nonlinear components or distortion to the signal. When the input signal possesses peak amplitudes which cause the amplifier to compress, to saturate (no appreciable increase in output amplitude with an increase in input amplitude) or to shut-off (no appreciable decrease in output amplitude with a decrease in input amplitude), the amplifier is being overdriven, and the output signal is clipped or distorted in a nonlinear fashion. Generally, an amplifier is characterized as having a clipping threshold, and input signals having amplitudes beyond the clipping threshold are clipped at the amplifier output. In addition to distorting the signal, the clipping or nonlinear distortion of the input signal generates spectral regrowth or adjacent channel power (ACP) that can interfere with an adjacent frequency.
Various linearization methods are used to enable the use of more cost-effective and more power efficient amplifiers while maintaining an acceptable level of linearity. Feed-forward correction is routinely deployed in modern amplifiers to improve the linearity of the main amplifier with various input patterns. The essence of the feed-forward correction is to sample the main amplifier output, isolate the distortion components generated from the main amplifier on a feed forward path by canceling the main signal components from the feed forward path. The distortion components are provided to a linear correction amplifier on the feed forward path which amplifies the distortion components. The distortion components on the feed forward path are maintained at 180 degrees out of phase to the distortion components on the main signal path and are combined with the distortion components on the main signal path. As the combined distortion components are 180 degrees out of phase, the distortion components cancel without affecting the main signal, thus providing a linear signal at the feed forward amplifier output.
Another linearization technique involves splitting a signal to be amplified by separate amplifiers of the same gain and power performance, and the amplified signal components are constructively combined at the output while the distortion components are used to cancel each other. FIG. 1 shows an amplifier circuit for amplifying an input signal Sin to produce an amplified output signal Sout. The input signal Sin can include CDMA or TDMA modulated RF carrier signals having respective fundamental frequencies f1 and f2. Both frequencies or signal components f1 and f2 can lie within standard wireless frequency bands in the 800-960 MHz vicinity. The various signals are shown in a vectorial fashion to conveniently illustrate phase relationships between the same frequency components at various points within the circuit 10. Thus a vector pointing in an upwards direction represents a frequency component of the opposite phase as the same frequency component represented by a downwardly pointing vector.
Input signal Sin is applied to input port 12 of a first coupler or power splitter 14 which splits signal into signal S1 at a coupled path output port 16 and a signal S2 at a direct path output port 18. The coupler 14 is preferably a passive device which may be a conventional branch line coupler or Wilkinson type divider that splits input power unequally between the two output ports, preferably with higher power being provided at port 18. For example, the signal level of signal S2 may be 10 dB higher than that of the signal S1. In this embodiment, the frequency signal components f1 and f2 produced on the direct path port 18 are delayed by a 90 degree phase shift while the frequency signal components on the coupled path port 16 have a 0 degree phase shift or no phase shift. Signal S1 contains only the frequency signal components f1 and f2 and is applied to a first amplifier 20 (A1) where it is amplified to produce an amplified signal S3 at the amplifier output. The amplifier 20 (A1) can be a conventional high frequency amplifier operating in class A, AB or B with power gain on the order of 30 dB to produce RF output power of 50 Watts, for example.
As is well known in the art, when a dual or multi-tone signal is applied to an amplifier, which is not perfectly linear, IMD products are generated at predictable frequencies. These IMD products are particularly apparent when the amplifier is being operated in saturation or in the gain compression region of the amplifier. The further into the gain compression region the amplifier is operated, the higher will be the IMD product levels. In addition, amplifiers which operate in class AB or class B modes tend to produce high IMD product levels when multi-frequency input signals are amplified. IMD product levels on the order of xe2x88x9230 dBc (30 decibels below the fundamental frequency or carrier level) are typical. Amplified signal S3 contains amplified frequency signal components f1 and f2 as well as undesirable intermodulation distortion (IMD) products or distortion components at frequencies f3 and f4, where f3 is typically a lower frequency than f1 and f4 is a higher frequency than f2. The frequency signal components f1 and f2 of the signal S3 are designated as having a zero degree phase shift, and the distortion components f3 and f4 are also designated as having a zero degree phase shift in brackets.
The amplified signal S3 is applied to input port 22 of coupler 24, which may be a conventional hybrid (e.g. branch line), backward firing or parallel-line coupler with a coupling value C22. In this case, coupled path signal S4 on output coupling port 26 will be 30 dB below the level of the direct path signal S8 emanating from direct port 25. The voltage levels of the frequency components of the signal S4 are each C22 times the corresponding voltage levels of the S3 frequency components. The voltage levels of the S8 are the square root of {square root over (1xe2x88x92C222)} times the corresponding voltage levels of the S3 frequency components. The phases of the frequency components of S4 will be equal to the corresponding ones of S8. When a branch line or other hybrid coupler is used for the coupler 24, then the signals S4 and S8 will differ in phase by 90 degrees. In this embodiment, the coupler 24 produces the signal S8 from the direct port 25 with the frequency components f1-f4 phase shifted by 90 degrees as designated by the xe2x88x9290 degrees and the xe2x88x9290 degrees in brackets. The phases of the frequency components f1-f4 of coupled path signal S4 from the coupling port 26 remain at 0 degrees.
Coupled path signal S4 is then applied to an attenuator 27 and a phase shifter 28. The attenuator 27 and the phase shifter 28 are designed to adjust the amplitude and phase of the signal S4 at each of the frequencies f1-f4. The amplitude of the frequency signal components f1 and f2 of the resulting signal S5 is adjusted to be smaller in amplitude than the frequency signal components f1 and f2 of a signal S6. The signal S6 flows into coupler 29 and is essentially signal S2 delayed by delay line DelayA. The phase shifter 28 adjusts the phase of the frequency components f1-f4 to be 180 degrees out of phase with the corresponding frequency components of S6 within the coupler 29. The coupler 29 receives the signals S5 and S6 and is designed to combine the signal S5 and the signal S6 to provide signal S7 which has frequency components f1-f4. Due to the combination of S5 and S6, the voltage levels of the f1 and f2 signal components of the signal S7 will be equal in amplitude to the f1 and f2 signal components of the signal S1 at the input to the first amplifier 20, and the frequency distortion components f3 and f4 are 180 degrees out of phase with the resulting frequency signal components f1 and f2. In this example, the coupler 29 provides a 90 degree phase shift to the frequency signal components f1 and f2 of S6 and combines the frequency signal components f1 and f2 with a cumulative 180 degree phase shift with the frequency components f1-f4 of the signal S5 with a 0 degree phase to produce the signal S7. The signal S7 has the f1 and f2 signal components with phase values (xe2x88x92180 degrees) equal to the phase values of the respective f1 and f2 signal components of signal S6 (xe2x88x9290 degrees) delayed by 90 degrees (xe2x88x9290xe2x88x9290=xe2x88x92180 degrees), and the frequency distortion components f3 and f4 have phase values (0 degrees) equal to the phase value of the respective f3 and f4 distortion components of signal S5 (0 degrees).
The signal components f1 and f2 of the signal S7 have phase values of xe2x88x92180 degrees, and the distortion components f3 and f4 of the signal S7 have phase values of 0 degrees. The corresponding frequency signal components of signal S8 produced from the direct path port 25 have phase values of xe2x88x9290 degrees, as well as the corresponding frequency distortion components of S8 which also have phase values of xe2x88x9290 degrees. The signal S7 is applied to an attenuator 32 and a phase shifter 34. The attenuator 32 and the phase shifter 34 are designed to adjust the amplitude and phase of the signal S7 of the frequencies f1-f4 prior to being amplified by a second amplifier (A2) 36. In this embodiment, the amplitude of the frequency signal components f1 and f2 of the signal S7 corresponds to the amplitude of the f1 and f2 signal components of the signal S1. The amplifier 36 amplifies the signal S7 to produce a signal S9 with a gain corresponding to the gain provided by the amplifier 20. In amplifying the signal S7, the amplifier 36 produces IMD products at the frequencies f3 and f4 which are in phase with the signal components f1 and f2. As such, the signal components f1 and f2 are amplified with phase values at xe2x88x92180 degrees producing distortion components f3 and f4 also with phase values of xe2x88x92180 degrees. The amplifier 36 also amplifies the distortion components f3 and f4 of the signal S7 which have phase values of 0 degrees. Because the amplified and newly produced distortion components at f3 and f4 are 180 degrees out of phase, the distortion components combine to reduce the amplitude of the distortion components at f3 and f4. In this embodiment, the distortion components with phase values at 0 degrees remain along with the amplified signal components f1 and f2 as the signal S9.
The signal S9 with signal components f1 and f2 with phase values at xe2x88x92180 degrees and distortion components f3 and f4 at 0 degrees is provided to a coupler 38 which combines the signal S9 with a signal S10 to produce the output signal Sout. The signal S10 is essentially the signal S8 delayed by delay line (DelayB) 40 with frequency components at f1-f4 which have phase values of xe2x88x9290 degrees. In this example, the coupler 38 delays the phase of the frequency components f1-f4 of the signal S10 by 90 degrees and combines the signal S10 with the signal S9. The signal components f1 and f2 of the signals S9 and S10 constructively combine because the signal components at f1 and f2 combine in phase, for example with phase values at xe2x88x92180 degrees. The distortion components f3 and f4 of the signals S9 and S10 destructively combine to reduce the amplitude of the distortion components at f3 and f4 because the distortion components f3 and f4 of S9 (with phase values at 0 degrees) are 180 degrees out of phase with the corresponding distortion components of S10 (with phase values at xe2x88x92180 degrees).
To provide improved combination of the signal S9 with a signal S10 at the coupler 38, the amplitude of the frequency components f1-f4 of the signal S7 are adjusted prior to the amplifier 36 such that the amplitudes of the frequency components f1-f4 of the signal S9 are substantially equal to the amplitude of the corresponding components of the signal S10. The attenuator 32 can adjust the amplitude of the frequency components f1-f4 of the signal S7 prior to the amplifier 36. The phase shifter 34 adjusts the phase of the frequency components f1-f4 of the signal S7 such that the signal components f1 and f2 of the signal S9 are in phase with the corresponding signal components of the signal S10 within the coupler 38. In doing so, the distortion components f3 and f4 of the signal S9 should be 180 degrees out of phase with the corresponding distortion components of the signal S10 within the coupler 38 such that the distortion components f3 and f4 are reduced.
To provide acceptable operation of the above amplifier architecture, the attenuation and phase control variables must be maintained in balance. An alignment procedure is necessary to be employed in the lab or production line to find the proper setting for each variable. The resulting settings are saved and stored in an on-board memory or look-up table and used to set the phase and attenuation control variables for proper operation. The amplifier architecture can have difficulty in achieving linearization because the adjustments in the phase and/or amplitude are performed on both the signal components f1 and f2 and the distortion components f3 and f4. A conflict can arise because the amplitude and/or phase settings for improved cancellation of the distortion components is not necessarily the amplitude and/or phase settings for the improved combination of the signal components.
The present invention involves independently adjusting the relative phase and/or amplitude between the signal components and/or the relative phase and/or amplitude between the distortion components to improve the combination of corresponding components. For example, a signal amplification system has first and second amplifier paths carrying replicas of signal components. On the first amplifier path, a first amplifier amplifies signal components and generates distortion components. A replica of the amplified signal components and distortion is provided to a coupling path. On the coupling path, the distortion components are isolated by canceling the signal components, and the distortion components are then amplitude and/or phase adjusted without a corresponding adjustment to the phase and/or amplitude of the signal components. The adjusted distortion components are coupled onto the second path where the signal components and the adjusted distortion components are amplified by a second amplifier. The amplified signal components and distortion on the second path are combined with the amplified signal components and distortion on the first path to constructively combine the signal components and destructively combine the distortion components. By independently adjusting the distortion components relative to the signal components, the phase and/or gain adjustments to the distortion and/or signal components can be made which improve the constructive combination of the signal components and/or the destructive combination of the distortion components.